July 1947 QST
Wax nostalgic about and learn from the history of early electronics. See articles
QST, published December 1915 - present. All copyrights hereby acknowledged.
Narrow-band frequency modulation (NFM) was a relatively new
technology in 1947, having been advanced significantly during
World War II. Amateur radio operators were just getting
their gear back on the air after having been prohibited from
transmitting for the duration of the war
Comes," January 1942 QST). Few were probably
thinking about adopting and exploiting new modulation techniques,
but for those who were and recognized FM as the path to the
future of radio, QST published this fairly comprehensive treatment
of both frequency modulation (FM) and phase modulation (PM).
Mathematically, FM is the time derivative of PM. Both modulation
schemes vary the carrier frequency in some proportion to the
baseband signal. Author Byron Goodman provides some insight
into the techniques.
Low-Frequency N.F.M. and the Differences Between Frequency
and Phase Modulation
By Byron Goodman,* W1DX
F.m. and p.m. seem to be surrounded by more mystery than
a.m. ever was. With experimental bands soon to be available
at 3.9 and 14.2 Mc., interest in the subject should be increased
considerably, so the following article is intended to clear
up some of the hazy points. A simple p.m. modulator is described
for those who want to get in on the ground floor.
Now that narrow-band frequency (and phase) modulation may
soon be permitted in portions of the 3.9- and 14-Mc. bands,
this seems like a good time to look at some of the points that
have not been covered in recent issues of QST. There seem to
exist several popular misconceptions of just what the modulated
carrier of an f.m. or p.m. signal looks like and acts like,
and it is the purpose of this article to attempt to dispel these
ideas and replace them with clearer pictures.
Everyone is familiar with pure amplitude modulation - a simple
picture of the distribution of energy in the spectrum of an
a.m. signal is shown in Fig. 1. The carrier frequency is represented
by a single vertical line, as in 1-A. If the carrier is modulated
by a single frequency, F, of sufficient amplitude to produce
100% modulation, frequencies called "sidebands" are developed
on either side of the carrier, with an amplitude equal to one-half
the carrier amplitude as in Fig. 1-B. They are spaced in frequency
from the carrier by the amount in cycles equal to the modulating
frequency, as indicated by + F and - F, relating to the carrier
frequency. If the modulating power is made up of a complex wave
that can be resolved into two frequencies, as in Fig. 1-C, sidebands
occur for each of the two components of the modulating frequency.
Speech is a complex form that is practically always made up
of two or more frequencies. However, the important thing to
remember about a.m. is that for each modulation frequency, there
exists a single pair of corresponding sidebands, and no more.
The simple representation in Fig. 1-C does not necessarily take
into account the relative phase of the modulating-frequency
components, but it is adequate to consider that the sideband
amplitudes never exceed half the carrier amplitude for 100%
modulation. Further, when a sideband on one side of the carrier
is at a maximum, the corresponding sideband on 'the other side
is also at maximum.
A simple phase-modulator unit that can be used to drive the
average crystal-oscillator stage. Receiving tubes are used throughout,
and the output is about one watt. The black shield houses one
of the coils. Power plug and gain control are at the rear.
Frequency and phase modulation do not lend themselves to
such simple pictures. It is generally understood that frequency
modulation is obtained by changing the carrier frequency at
the frequency of the modulation, and the greater the amplitude
of the modulation the greater the frequency change, or deviation,
from the mean carrier-frequency. Phase modulation, on the other
hand, is obtained by shifting the phase of the carrier frequency
at the modulation frequency, and the greater the amplitude of
the modulation the greater the phase change. With a little thought
it can be seen that as the phase of the carrier is changed,
by speeding up or slowing down the r.f. alternations during
the audio modulation cycle, the frequency of the carrier must
change at the same time, since more or fewer alternations than
normal must occur during the speeding-up or slowing-down process.
Hence f.m. and p.m. are similar in that the carrier frequency
is changed during the modulation cycle.
Then it starts to get complicated! When a single tone is
used to modulate a carrier in either phase or frequency, not
a single pair of sidebands results, as with a.m., but theoretically
an infinite number of sidebands develops. The magnitude of the
sidebands depends upon the amplitude of modulation, with the
sidebands close to the carrier being the larger and the remote
sidebands existing only theoretically for all practical
purposes. To examine the procedure in an orderly fashion,
assume a carrier modulated by a single 1000-cycle tone. With
very little modulation, only the first pair of sidebands have
any significant amplitude, and the picture looks similar to
the one for low-percentage amplitude modulation. As the degree
of modulation is increased, the second and third and higher-order
sidebands begin to be significant, as shown in Figs. 2-B, 2-C
and 2-D. Notice that the sidebands occur at regular 1000-cycle
intervals. This is a significant point, since many seem to believe
that f.m. (or p.m.), can be made to occupy less spectrum space
than a.m. by keeping the frequency swing low. Such is not the
case - the instant any modulation is applied to the carrier,
sidebands exist removed from the carrier by the modulation frequency.
Fig. 3. shows how the amplitude of the sidebands varies with
the index of modulation - for some degrees of modulation, some
sidebands disappear and so does the carrier. Also, some pairs
of sidebands will be out of phase with each other for high degrees
Fig. 1 - Spectrum analysis of an a.m. signal.
The unmodulated carrier is shown at A, modulation by a single
audio frequency is shown at B, and modulation by two frequencies
is represented in C. There is a single pair of sidebands for
each modulating frequency, and the sidebands are removed from
the carrier by that frequency.
What we have elected to call "degree of modulation" in the
above discussion is more correctly known as the" modulation
index." The modulation index is defined, for f.m., as the ratio
where Δƒ is the frequency deviation and F the highest
modulation frequency. For p.m., it is defined simply as the
phase change, in radians (one radian = 57.3 degrees). The curves
in Fig. 3 apply to either f.m. or p.m., if the modulation index
as described above is substituted for "degree of modulation."
The definition of modulation index helps to clarify the distinction
between f.m. and p.m. and how their audio characteristics differ
when rectified by the same detector system. In an f.m. system,
for example, a modulation index of 2.0 means that the maximum
deviation divided by the highest modulation frequency is equal
to 2.0. If the top audio frequency is, for example, 5000 cycles,
then the maximum deviation will be ± 10 kc. The f.m.
transmitter with an index of 2.0 and a top audio frequency of
5000 cycles will deviate ± 10 kc. for full modulation
at any audio frequency below 5000 cycles. Any greater deviation
doesn't mean "overmodulation" as we know it for a.m., but simply
that the signal can no longer be described as having an index
of 2.0. If the detection system is designed to give maximum
output for a deviation of ± 10 kc., then a greater deviation
will result in distortion in this detector, and it might be
described as "overmodulation," but only for that particular
detector. Whether the modulating signal is 100 or 5000 cycles,
maximum undistorted output from the detector will be obtained
when the deviation is ± 10 kc. in either case. Note that
this represents a modulation index at 100 cycles of 50. If Fig.
3 were extended to values of index ("degree of modulation" in
sketch) of 50, it would be found that the total number of significant
sidebands would be multiplied enormously. But with the modulation
frequency of 100 cycles, these sidebands are now only 100 cycles
apart, and actually the significant sidebands (less than 30
db. down) do not extend out as far as the fewer but more widely
separated sidebands of the 5000-cycle modulation frequency.
It is therefore readily apparent from the examples in this discussion
that, with f.m., the modulation index gives rather incomplete
information on the bandwidth unless the highest audio frequency
is also specified.
Fig. 2 - Spectrum analysis of an f.m. or
p.m. signal. The unmodulated carrier is shown at A, and modulation
by a single frequency is shown in B, C, and D. B corresponds
,to a low degree of modulation, C and D show what takes place
as the modulation is increased. All sidebands are spaced an
amount equal to the modulating frequency. Note that the amplitude
of the carrier decreases as the modulation is increased.
In p.m. and any given modulation index, the number and amplitude
of the sidebands are exactly the same for any single modulation
frequency, since the index in this case is only the number of
radians of phase swing either side of zero required to give
the phase modulation. However, the sidebands are separated by
the modulation frequency, so a low frequency of modulation will
result in a narrow bandwidth and a higher modulating frequency
will cause the signal to occupy more spectrum space. If the
index is low (0.5 or less), so that only the first sidebands
can be considered as significant, this gives a spectrum picture
quite similar to that for a.m. Fig. 4 illustrates the comparative
spectrum space occupied by f.m. and p.m. signals for different
modulation indices and modulation frequencies. The p.m. pictures
show immediately why pure p.m. received on an f.m. detector
will be lacking in "lows," since the f.m. detector requires
that modulations of equal intensities but different frequencies
have roughly the same deviations. The condition can be corrected,
of course, by attenuation of the "highs" at the transmitter
or receiver. The latter is generally more convenient, since
most receivers have a "tone" control that can be cranked over
to give the necessary attenuation of the higher frequencies.
One point worth noting about p.m. is the fact that it does
not necessarily require a long string of frequency multipliers
following it in order to obtain a usable index. Readers familiar
with the Armstrong system of f.m. know that p.m. is used at
a low level and converted to f.m. Such a system does require
considerable multiplication, for reasons that will be described,
but this is only because f.m. is the desired end product. It
is not impossible to obtain an index of 0.5 in p.m. without
any multiplication, and, indeed, this seems to hold the most
promise for simple work on the 75-meter band.
Reference to Fig. 5 will show why f.m. obtained by phase
modulation requires so much multiplication. For a p.m. signal,
plotting deviation vs. modulating frequency gives the solid
line sloping up from the origin. Since f.m. requires nearly
the same deviation for all modulating frequencies, it is necessary
to modify the audio characteristic, as shown by the dotted line.
This results in the f.m. characteristic indicated by the dashed
line, but note that it reduces the deviation to that obtainable
through p.m. at the lowest usable audio frequency. To minimize
distortion, the phase modulation is held down to a low level
anyway, and the total result is a very low modulation index
at the control frequency. Such technique is not necessary in
amateur work, and hence p.m. looks good for our low-frequency
bands. Phase modulation suffers in its ability to reject noise,
but this is not under discussion, although it is of course a
big point in the Armstrong system, and accounts for the use
of f.m. and a high index.
Fig. 3 - Showing how the amplitude of the
sidebands of an f.m. or p.m. signal varies as the modulation
is increased. If the curves were extended for greater values
of "degree of modulation," it would be seen that the carrier
value goes through zero at several points, as do the various
sidebands. Amateur n.f.m. and n.p.m. should be confined to a
degree of modulation equal to 0.5 or 0.6, so the additional
sidebands are not significant.
It is important that one more point be clarified in this
discussion. The pictures given for single-tone f.m. or p.m.
show a rather discouraging bandwidth for any narrow-band application,
if exactly the same condition were to hold for complex modulation
by two or more frequencies. However, when a complex wave is
used to frequency - or phase-modulate a carrier, the resulting
sidebands are not the same as would be obtained by superimposing
the pictures of modulation by the component tones. The existence
of the sidebands in f.m. or p.m. always results in the reduction
of the carrier-frequency amplitude (see Figs. 2 and 3), and
the total energy always remains the same. If two sets of sidebands
exist, corresponding to two modulation frequencies, both of
these sets of sidebands draw from the carrier, and the resultant
effect is to reduce the amplitude of the sidebands, since the
carrier sets the limit on total available energy. For this reason,
a single audio frequency will yield higher-order sidebands than
will a complex wave of the same amplitude. This means that a
single tone applied to an f.m. or p.m. signal may show several
sets of sidebands, while voice modulation of the same amplitude
will not show as many. As a result, speech occupies less channel
space than a single tone of the top frequency existing in the
speech for a given modulation index.2
Fig. 4 - A comparison of the bandwidths of
f.m. and p.m. signals for different modulation frequencies.
A modulation index of 1.0 and an upper frequency limit of 4000
cycles are assumed. Note that the sketch for f.m. with an index
of 10 (lower left) shows the relative phase of the sidebands.
Bandwidths are based on neglecting sidebands down more than
Another point that is often confusing is what happens to
f.m. and p.m. signals at the harmonic frequencies of the carrier.
At first glance, one might think that, if a carrier is frequency-modulated
at its fundamental by a 1000-cycle tone, to give a pair of sidebands
removed from the carrier by 1000 cycles, then the carrier and
the sidebands would have harmonics, and so the sidebands would
move out from the carrier at the harmonic frequencies. However,
this is not the case, any more than it is with a.m. Unfortunately,
there is no simple physical picture that can be given of the
process of modulation of any type. We all know that sidebands
are generated under modulation, and it can be shown readily
by mathematics that the sidebands will appear. We can show the
existence of the sidebands with a "spectrum analyzer," but they
just seem to be something we have to accept on the basis of
mathematical and practical proof. It isn't too many years since
the "great sideband controversy" raged between the English and
American engineers, the English holding that sidebands existed
only in the mathematics.
Fig. 5 - The necessary audio correction (dotted
line) to correct a p.m. characteristic (solid line) to f.m.
(dash line). Note that this limits the resultant f.m. deviation
to the highest uncorrected p.m, deviation. This is the principle
used in wide-band f.m. transmitters, but it is not necessary
for amateur work.
The harmonics of the carrier result from distortion of the
carrier in some nonlinear element, such as a vacuum tube. But
the sidebands are a result of the operation performed on the
carrier (a change in amplitude, frequency or phase). The change
of frequency or phase is multiplied in direct proportion to
the frequency multiplication generating the carrier harmonic
frequency, but the sidebands set up are the same as those produced
by direct modulation of a carrier fundamental but with the greater
index of modulation. Hence in most f.m. and p.m. work it is
customary to do the modulation at a low frequency and frequency-multiply
until the necessary index is obtained.
It will be necessary for any operator using narrow-band f.m.
or p.m. to check his transmissions and to be sure that his signal
is occupying no more spectrum space than an a.m. signal, in
keeping with the definition of n.f.m. The future of f.m. and
p.m. in the amateur bands depends on those who use it during
the trial period and, since it is proving to be such a boon
to those heckled by BCI, it would be unwise and unfair to jeopardize
its future by giving it a bad name on the air - and with the
FCC monitoring stations. For this reason, it is the duty of
every user of f.m. and p.m. to do his best to insure that his
equipment is properly checked and monitored. Unfortunately,
this is not a simple problem, and no such clean-cut solutions
as exist for a.m. are known at the present time.
In the case of wide-band f.m., it is possible to apply single-tone
modulation and increase the modulation until the carrier disappears.3
This corresponds to an index of 2.4 for the first disappearance
of the carrier, and 5.5 for the second. This is not a convenient
method, however, for a station operating on 14 or 28 Mc., unless
the operator has access to stable receiving equipment at 56
Mc. or some such high harmonic frequency. One would then establish
the audio level required to give the necessary deviation, knowing
that the index would be divided by the order of multiplication,
and then keep his modulation below this level, by means of a
'scope or some other audio-level indicator. Figuring on an operating
index of 0.6 at the operating frequency, this method would require
a 56-Mc. receiver to check a 14-Mc. signal and a 112-Mc. receiver
to check a 28-Mc. n.f.m. transmission, either one with excellent
frequency stability and good selectivity. A 3.9-Mc. signal can
be checked, of course, by a receiver capable of tuning to 15.6
Another method, which is unfortunately beyond the reach of
most amateurs, is to use a special "spectrum analyzer" designed
This is an instrument similar in principle to a panoramic
receiver, but in this case it requires a few refinements such
as a crystal-filter i.f., a slow sweep rate, and a long-persistence
screen (to handle the slow sweep rate). Such a device would
make an excellent club project, but it is hardly likely to become
standard ham-shack gear. However, it is quite possible that
it will be the sort of thing the FCC monitoring stations will
use for checking amateur n.f.m. transmissions, since it offers
an instantaneous picture of the bandwidth. This is no hardship
on the n.f.m. gang, however, because the same instrument turned
on maladjusted a.m, signals would also tell the sad story about
Since the precise methods are involved and expensive, we
must fall back on something a little more simple and less complicated.
One redeeming feature of n.f.m. is that one can listen to it
as it will sound on the air without turning on the whole transmitter,
and this provides an excellent opportunity to do all the testing
without putting the signal into an antenna. By tuning in the
harmonic of the n.f.m. unit on the band where operation is to
take place, one can get a rough check on the bandwidth by noting
how much room the signal takes in comparison with a.m. signals
on the same band. The most precise method available to the average
amateur to measure the bandwidth is to use his communications
receiver and its crystal filter as a sort of "poor man's spectrum
analyzer." If the receiver is accurately calibrated in kilocycle
steps - which means that the absolute calibration can be off
but the kilocycle divisions accurate - or if a suitable calibration
chart can be made to find out how many dial divisions per kilocycle
exist at the operating frequency, the problem is fairly simple.
The receiver must be stable, of course, and any measurements
should only be made after a suitable warm-up period. Suppose,
for example, that one is setting up his f.m. or p.m. signal
in the 14-Mc. band. The oscillator and possibly one or two following
stages are turned on, to give a reasonable signal in the receiver
tuned to 20 meters. The signal level should be equivalent to
an average signal in the band, as judged by the S-meter, and
it may be necessary to short the input of the receiver to get
it, unless the oscillator and other stages are operating at
a low power level. The receiver crystal filter is set to its
sharpest position, the b.f.o. is turned on and the a.v.c. turned
off. It may be necessary to reduce the r.f. gain slightly, to
avoid overload of the receiver. Tune the carrier on the peak
of the crystal and set the b.f.o. for the usual beat note, around
500 or 600 cycles. If the receiver is accurately calibrated,
either by the manufacturer or by the operator making a calibration
curve, detune it exactly 3 kc. If the calibration isn't available,
it will be necessary to modulate the f.m. or p.m. unit with
a 3000-cycle tone, which can be obtained from an audio oscillator
built or borrowed for the occasion. The 3000-cycle modulation
will cause sidebands to appear on either side of the carrier,
spaced by 3000 cycles, and one of these will serve as a reference
point. The amplitude should be kept low, so as not to introduce
more than one pair of detectable sidebands.
After the receiver setting is established 3 kc. off the carrier
frequency, talking into the microphone and experimenting with
various voice levels will give some level at which the voice
is heard to splash over occasionally. This represents the upper
limit of modulation level that should be used. Once the proper
level has been established, it can be monitored by a 'scope
or other voice-level indicator connected in the audio amplifier
ahead of the frequency or phase modulator, unless one is willing
to run the risk of depending upon the setting of the gain control
and one's memory of his voice level, bearing in mind his responsibility
not to give n.f.m. a bad name, or the FCC a chance to tag him.
There is nothing simple that can indicate directly from the
carrier, as in a.m. work, since a properly-adjusted f.m. or
p.m. transmitter will be accompanied by no amplitude changes
If the operator's voice is naturally high-pitched, the 3-kc.
figure may be slightly unfair, and perhaps 3.5 or even 4 kc.
is a more reasonable limit. However, there aren't very many
necessary components existing in normal speech that run this
high, if there are any, and they are just as likely to be introduced
by distortion in the audio amplifier or modulator. The best
practice, as in a.m., is probably to limit the upper response
of the audio amplifier to the useful frequencies below 3 or
3.5 kc., by means of suitable filters.
A Simple Phase Modulator
During the past few years, a number of different types of
phase modulators have been described in the literature. The
new Raytheon cascade-modulation system is interesting, but it
requires a number of stages and the tuning procedure does not
lend itself too well to rapid frequency change, as is often
required in amateur work. Other systems using balanced modulators
in one form or another are at a disadvantage mainly because
their apparent complexity will frighten a few potential customers
for p.m., although they are actually quite satisfactory in every
One of the attractive things about p.m. is that it can be
applied to the transmitter at some point other than the oscillator,
without any alterations that might impair the frequency stability.
Until we have quite accurate methods for measuring and insuring
bandwidth of f.m. transmissions, it seems highly desirable to
avoid the use of direct reactance modulation on the oscillator
to obtain an f.m. signal. This fact alone makes p.m. a natural
for amateur use. Also, it is more difficult to obtain a high
index of modulation with p.m. than with f.m., so the bandwidth
is inherently more limited.
Fig. 6 - A simple phase-modulator unit.
The simplest phase modulator we have been able to find is
one suggested in Mr. Hund's book.5 The author suggests
using a reactance modulator across the tuned plate circuit of
a driven r.f. pentode amplifier. When the reactance modulator
changes the tuning of the circuit in accordance with the modulation,
the phase angle of the effective tuned circuit is changed and
hence the phase of the voltage developed across it. Since the
phase change across a tuned circuit of Q = 10 or higher is fairly
linear for a range of ±25°, all that is required is a reactance
modulator capable of detuning the tuned circuit the necessary
amount. Assuming a Q of 20 for the circuit, an angle of 26.5°
is obtained when the detuning is an amount equal to
This works out to be
= 0.04875 Mc. = 50 kc. approx. A Q of 20 will be obtained at
3.9 Mc. with a total tank capacity of 50 μμfd. and an
effective parallel resistance of 16,300 ohms (from Q = 2πƒRC).
A change of ± 1.2 μμfd. will swing the 50-μμfd.
tank ± 50 kc., and this is easy to obtain with a reactance
modulator. From the design equations,6 this can be
obtained with a reactance modulator using an inductive element
of 2.5 mh., a resistive element of 0.5 megohm and a mutual conductance
change of ± 240 μmhos. This is a reasonable range
for almost any of the better receiving-type pentodes.
An experimental model was built and is shown in the photographs.
The wiring diagram, shown in Fig. 6, shows how simple the unit
can be. A 6SJ7 speech amplifier builds up the signal from a
crystal microphone sufficiently to give enough swing for the
reactance modulator. A gain control, R5, allows the
gain to be reduced when the transmitter output is on 14 or 28
Mc., since the multiplied modulation index at these frequencies
might be too high. The reactance modulator is slightly different
than those previously described in that it uses an inductance-resistance
divider, RFC1R6, to obtain the quadrature
current rather than the more usual condenser-resistor combination.
The principle, however, is practically the same, and it requires
no elaboration here.
A Tri-tet oscillator is used, with straight-through operation;
i.e., the plate circuit is tuned to the crystal frequency. Since
this type of operation requires a well-screened tube, the 6SK7
was selected. The effect is the same as if a separate crystal-oscillator
tube were used to drive an amplifier, since the plate-circuit
tuning or loading has no effect on the crystal oscillation.
This is important if one is to obtain pure phase modulation.
If VFO were to be used, the VFO would feed into a tuned circuit
between grid and ground of the 6SK7, and the tuned cathode circuit
would be replaced by a bias resistor and by-pass condenser.
In the unit shown, the tuned cathode circuit is resonant around
4.5 Mc. Its tuning will affect the amount of oscillator output
slightly, but the major control of output is the value of oscillator
screen voltage. This was made convenient to adjust in the model
by bringing out the lead separately (marked "screen ") and running
it to the regulated 150 volts through an adjustable resistor.
The value isn't critical, and several fixed resistors are all
that is necessary to make the adjustment. The oscillator output
must be adjusted to avoid overdriving the amplifier. The inductance
L2 is shielded to avoid self-oscillation in the amplifier,
and the plate by-pass condenser, C16, is mounted
across the tube socket to shield the grid and plate pins from
Since this particular unit is only a model and will probably
not fit too well into anyone's ideas about how such units should
be constructed, only the tuning details will be included. The
operator with VFO can use the circuit by making the oscillator
changes mentioned earlier, and the station requiring more power
output from the unit will require additional power stages following
the 6SG7 amplifier. The output of this little unit is enough
to light a small pilot lamp, representing about one watt of
power, enough to drive the usual crystal-oscillator stage. The
direct substitution of larger tubes throughout the unit is not
recommended, unless a well-shielded tube like the 802 is used,
since one is likely to encounter the usual difficulties with
feed-back if beam tetrodes are used.
The first step in putting the unit in operation is to adjust
the crystal oscillator. With the screen of the oscillator connected
directly to the 150-volt source, and with normal voltages on
the rest of the unit, adjust the cathode-circuit condenser,
C9, until the crystal oscillates. A 0-1 milliammeter
between the bottom of R11 and ground will serve as
an output indicator, and a receiver should be used as an additional
check on the signal. When oscillation of the crystal has been
checked, add resistance in the oscillator screen lead until
the grid-current reading reaches a low value, of around 0.1
ma. or less. It should still be possible to swing the tuning
of C11 without throwing the crystal out of oscillation
or even affecting the frequency. If it can be thrown out of
oscillation, readjust C9 or reduce the value of the
screen-dropping resistor. In the unit shown, 0.2 megohm could
be connected in the screen lead without stopping crystal oscillation.
If a VFO is being fed into the unit, the screen voltage of
the 6SK7 should be reduced until the drive on the 6SG7 amplifier
is as specified for crystal operation.
Using a small lamp load or the grid current of the stage
the 6SG7 is driving, resonate the output circuit L3C17.
If it tunes broadly, it probably indicates that the stage is
being overdriven, or that the 6SG7 is oscillating, although
no trouble with oscillation was encountered in this unit. The
modulated circuit, L2C11, will tune broadly
because it is loaded by R11, but it should be centered
on the broad resonance peak or otherwise the modulation will
Talking into the microphone and monitoring the signal on
14 Mc. will give you a check on the modulation, in the manner
described earlier in this article. It will be found that more
than enough modulation can be obtained for 14 Mc. when using
an 80-meter crystal, but on 3.9 Mc. the best reception is obtained
when using crystalfilter reception methods, as outlined previously.7
For 3.9-Mc. work, it would probably be better to do the modulating
at 1.95 Mc.
No great claims are made for the unit, except that it is
a simple thing to get going and it will enable all of us that
are interested to take advantage of the opening of the lower-frequency
bands to p.m. Somewhat greater swing can be obtained by increasing
the value of R11, and this might be necessary if
a low-output microphone is used. If listeners report "no lows,"
explain that you're using p.m. and suggest that they crank up
the tone control on their receivers. However, a cheap crystal
microphone may have poor low-frequency response, so the fault
may be in your own equipment if you are using a bargain microphone.
Good practice would indicate a low-pass filter ahead of the
reactance modulator, to limit the high-frequency response and
consequently, the bandwidth, and such a filter could be put
in the circuit ahead of the gain control.
A view under the chassis of the phase modulator.
The Tritet cathode circuit is mounted on the side of the chassis
near the microphone connector.
*Assistant Technical Editor, QST.
1 The information for these sketches and
for Fig. 2 was obtained from Hund's Frequency Modulation, McGraw-Hill
Book Company, an excellent text for further study of the subject.
2 Crosby, "Carrier and Side-Frequency Relations
with Multi-Tone Frequency or Phase Modulation," RCA Review,
3 Crosby. "A Method of Measuring Frequency
Deviation." RCA Review, April, 1940; also: Grammer, "Getting
on 56-Mc. F.M.," QST, June, 1940.
4 Marks, "Cascade Phase-Shift Modulator,"
Electronibs, December, 1946.
5 See Footnote 1.
6 Hund, Frequency Modulation, p. 166.
7 Grammer, "N.F.M. Reception," QST March
Posted June 11, 2015