September 1973 Popular Electronics
Table of Contents
Wax nostalgic about and learn from the history of early electronics. See articles
published October 1954 - April 1985. All copyrights are hereby acknowledged.
Mr. Lothar Stern, of Motorola
Semi, published a 3-part series on transistor theory in Popular Electronics magazine
in 1973. This is part 2.
Part 1 introduced
the basics of the bipolar transistor, and this follow-on article starts addressing transistor
circuit configurations - common emitter, common gate, common collector, Darlington, differential
- as well as presenting gain equations and delving a bit into the physical construction of
the semiconductor elements. I do not yet own the October 1973 issue with Part 3,
but when I do, I'll post it.
Do You Know Your Bipolar Transistors?
Part 2 of a 3-Part Series on Basic Transistor Theory
By Lothar Stern, Motorola Semiconductor Products Inc.
Fig. 5 - Conventional common-emitter bias circuits. Table gives approximate
Fig. 6 - Circuit of a typical common-emitter RC-coupled amplifier and its
ac and dc loading curves.
Fig. 7 - Equivalent high-frequency common-emitter circuit (a) and its response
Fig. 8 - One-stage amplifier and equations for feedback effects.
Fig. 9 - Darlington transistor pair.
Fig. 10 - Basic differential amplifier.
Fig. 11 - Typical alloy transistor.
Fig. 12 - The microalloy structure.
Fig. 13 - Microalloy diffused type.
Fig. 14 - Epitaxial mesa structure.
Biasing. When operated as an amplifier, the transistor must first be biased
to some quiescent value of collector current, so that both positive- and negative-going input
voltage excursions will cause corresponding changes in output voltage and current. The ideal
bias point is represented by Q on the loadline because this permits approximately equal excursions
in IC and VCE in both directions along the load line without signal
clipping. The bias point is established by a quiescent base current that results in a dc collector
current of approximately IC(sat)/2.
Several circuits are used for establishing the bias point. Among the most familiar are
those in Fig. 5. The basic performance difference is in the bias-point stability. At point
Q on the load line in Fig. 4, the transistor has a beta of approximately 20. If a transistor
with a beta of 40 were substituted (simulated by dividing all IB values by 2),
and if IB were held by the bias circuit to 2.5 mA, as before, the operating point
would move up the load line to point Q', a much higher value of IC. As a result,
considerable distortion would occur for high-value input signals.
The bias point stability factor (S) is defined as the percent-change in IC,
for a percent-change in β, or ΔIC/IC = SΔβ/β.
If a percent-change of β causes a corresponding percent-change in IC, the
least desirable condition, then S = 1. If IC is independent of β (corresponding
to a zero change in IC when β is varied), then S = O. The formulas accompanying
Fig. 5 give IC and S as functions of β and assign values for S under specific operating
conditions. The bias arrangements in Fig. 5c and 5d using emitter degeneration are preferred
because, by proper choice of resistor values, the effect of β on IC can be
made almost negligible. This prescribes a large value of RE, so that the voltage,
IERE, at the emitter is much larger than VBE or IBRB.
To prevent degenerative ac feedback, RE is normally bypassed by a large-value capacitor.
(Figure 5c is used when a positive and negative power supply is available. For single-supply
operation, Fig. 5d is preferred.
In practical transistor amplifiers (RC coupled amplifier, for example) the operating point
is influenced by both dc and ac conditions. Figure 6 shows a typical RC-coupled amplifier
and its representative loadline plot. Note that there are two loadlines - a dc loadline whose
slope is affected only by the value of RC and an ac loadline whose slope is determined
by rL, the equivalent resistance of RC and RL in parallel.
The dc loadline represents the path along which the operating point can be established. The
ac loadline intersects the dc loadline at the operating point, and the actual signal varies
along the ac loadline, which sets the V and I output limits.
The ac performance of the circuit in Fig. 6 can be established from the high-frequency
equivalent circuit in Fig. 7a. (For this approximation, it is assumed that the signal frequencies
are high enough that all capacitive reactances of Fig. 6 are negligibly small.)
Each transistor junction has an associated junction capacitance. These are quite small
(on the order of a few picofarads), but they do affect transistor action at high frequencies.
A typical transistor frequency response plot is shown in Fig. 7b. At the frequency where the
reactance of the parasitic input capacitance equals the input resistance, βre,
the current to the input resistance is bypassed through the capacitance to the point where
the effective β is down 3 dB from its low-frequency value. This is called the β-cutoff
frequency, fae. If the frequency is further increased, β continues
to decrease at a rate of 6 dB per octave. The frequency at which β equals unity is specified
on data sheets as fT, the current-gain/bandwidth product. Given fT,
it is possible to determine transistor β for any frequency between fae
and fT from the relation β = fT/f.
Negative Feedback. While the dc degenerative feedback associated with
RE of Fig. 6 stabilizes the operating point, making it independent of changes in beta and
other temperature-dependent parameters, the bypass capacitor keeps it from compensating for
the deleterious effects of these changes on the ac signal. Moreover, while proper placement
of the operating point can reduce non-symmetrical signal clipping, it cannot reduce the distortion
for large signal swings caused by nonlinearity of the ICIB, characteristics
(Fig. 4). These characteristics can be greatly improved by means of negative signal feedback,
which requires a small unbypassed resistor, rE, in series with RE as
shown in Fig. 8. (This is only one of many possible feedback arrangements.) In addition, negative
feedback improves frequency response and compensates for changes in output voltage (and gain)
due to variations in temperature-sensitive parameters such as re, and βac.
The equations accompanying Fig. 8 describe the basic advantages achieved through negative
feedback, as well as the price paid for them in terms of voltage gain. However, since feedback
increases input resistance, the loss of gain can partly be recovered because of an increase
in the gain of a previous stage caused by the increase in input resistance.
Darlington Transistors. Modern semiconduotor technology not only has led
to complete circuits on a single chip of silicon (integrated circuits) but also to compound-connected
transistors. For the circuit designer, the latter provides some cost and space savings, while
still permitting unrestricted circuit design freedom. One of these devices is the Darlington
pair shown in Fig. 9.
Though consisting of two interconnected transistors, the device can actually be treated
as a single transistor with extremely high current gain and input resistance. Normally, Darlington
pairs are employed in the grounded collector configuration. Commercially, they are available
as small-signal and power devices, in both npn and pnp polarities and with betas ranging from
several 100 to several 1000.
Differential Amplifiers. With the advent of integrated circuits, the circuit
in Fig. 10 has become increasingly important. Being de coupled through a common emitter resistor,
it has no low-frequency limit; but, unlike other types of dc-coupled amplifiers, it exhibits
excellent stability and drift-free operation without requiring elaborate compensating circuitry.
This is its most important characteristic. Operated in the differential mode, as shown, the
output voltage responds only to difference inputs to the two bases. If a common-mode signal
were applied (as in the case of ground line or power supply noise) or if the characteristics
of the transistors were to change in response to a change in temperature, the collector current
of both transistors would be affected equally. As a result, the output voltage between the
collectors would remain constant.
Transistor Fabrication Processes.
Over the years, many processes and structures have been used in transistor fabrication.
Most of them are still being used, though the older processes no longer offer the best obtainable
performance. The major sequential developments in the processing of the bipolar transistor
are shown in Figs. 11 through 15.
In Fig. 11A is a typical alloy transistor, while Fig. 11B shows its impurity profile. It
is simple and inexpensive to build. It provides excellent low-frequency beta and can operate
at high currents and power levels, but not at high frequencies or high voltages.
Figure 12 shows the construction detail and impurity profile of a typical microalloy (MAT)
structure. It is similar to the technique shown in Fig. 11 except that shallow pits are etched
into the base substrate prior to collector and emitter alloying. The thinner base improves
the frequency response but results in a fragile structure and further reduces breakdown voltage.
The process shown in Fig. 13 uses diffusion of impurities into a thin base membrane prior
to alloying to permit a closely controlled, graded impurity profile. This technique offers
frequency responses up to 100 MHz.
The process shown in Fig. 14, with extremely thin collector and base regions and unrestricted
use of different material resistivities, provides high-frequency performance up to a gigahertz.
It also provides high gain and high breakdown voltage. However, sensitive pn junctions are
exposed to the atmosphere, resulting in high leakage current.
Posted September 7, 2017